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Adventures in induction heating

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Wolfram

Fri Jan 16 2015, 05:08PM

Registered Member #33 Joined: Sat Feb 04 2006, 01:31PM
Location: Norway
Posts: 971

This thread is both to document my experiences and experiments in induction heating, and to present what I have learned along the way. Like so many hobby projects, this one has a very real chance of ending up as an unfinished pile of components, so by documenting what I've done so far, at least some use can come out of it. Along the way, a lot of my preconceptions about induction heating have been proven wrong, so by sharing what I've learned, I can hopefully dispel some common myths. My ultimate goal is to create a compact stand-alone induction heater circuit that can drive a variety of tank circuits, both LCLR and series resonant, at up to 500 kHz and possibly higher. It should be able to handle as much power as a regular european outlet can supply, that is around 20A at 230V, and the power level should be adjustable, with a power factor close to unity. Further, it should be robust enough to be used in a workshop setting for normal everyday tasks. Lastly, it should not use any exotic or expensive components, aside from the tank capacitor. From my research so far, I believe this is possible, but not trivial.

To not lose everybody's interest by starting with theory, I'll begin with the results of my experiments so far. To get a feeling of how everything works together, I made a simple open-loop LCLR half bridge induction heater. The tank circuit was construced from a Celem CP80/200 400nF 500V 400A mica capacitor and a 500nH workcoil made from 5mm copper pipe. An 8ÂµH matching inductor made from CAT5 network cable connected the tank circuit to a MOSFET half bridge. The MOSFETs were driven by a pair of IXDN614 gate drivers, through a gate drive transformer. An open loop oscillator made from the VCO section of a 74HC4046 provided the operating frequency. Power for the half-bridge was provided by an isolated 3A Variac, and a bridge rectifier. DC bus capacitors were kept minimal to ensure a good power factor. This has an added advantage of keeping the stored energy low, minimizing any damage in case of a MOSFET failure. Here is a video giving an overview of the test setup

So far, the circuit has performed well, and it has given me a good feel of how the induction heater responds to different load conditions and operating frequencies. The tank circuit resonates at around 360 kHz unloaded, and I've had it up to 400kHz when loaded with large conductive objects. I've mostly limited myself to the 3A that the variac is rated for, .but during some more enthusiastic moments I've had the 3A meter needle pegged, probably at somewhere north of 4A (at 250V AC). It melts thin pieces of steel without any problem, here is a video of it melting a bracket from a network card, which is made of 1mm thick steel

The MOSFETs I used had about 10nF of gate capacitance each, and at the frequencies I'm using it's difficult to get reasonable rise times with a GDT. Therefore, it has been difficult to introduce dead time to get ZVS. The MOSFETs also have significant conduction losses, even though they are some of the lowest Rdson 500V MOSFETs I could find. IGBTs would fix both these problems, so for the next prototype I'm going for IGBTs. The downside with IGBTs is tail current losses, which can be fixed by ensuring ZCS. The easiest way to do this is by switching them at the tank resonant frequency. This rules out power control by detuning, further complicating things.

The next post will be about my prototype mark II, or maybe some theory.

Wolfram

Mon Feb 02 2015, 11:23PM

Registered Member #33 Joined: Sat Feb 04 2006, 01:31PM
Location: Norway
Posts: 971

For Mk II of the prototype, I'm going with a transformer coupled series resonant tank circuit. Not neccessarily because I think it's superior to LCLR in any way, but I want to try it as a fair comparison.

The driver is completely revised. A 4046 PLL is used to control the switching, with a loop filter design stolen from Steve Conner's DRSSTC controller. Phase comparator 1 is used, due to its good immunity to noise. The PLL locks the phase angle between the VCO and the power capacitor voltage, with a selectable phase offset. This allows the switching phase angle to be adjusted between early and late, allowing both ZVS and ZCS.

The MOSFET halfbridge has now been replaced with an IGBT fullbridge using some new cheap Fairchild FGT40T65SPDs on a common heatsink, isolated with kapton tape. The heatsink is an old Intel Pentium 4 one with a small fan, it's a bit on the small side, but its small thermal mass makes it easy to get an idea of the IGBT power dissipation as a function of switching phase angle and loading. As mentioned in the previous post, the IGBTs have much lower input capacitance, 1.4 nF vs 10 nF, and conduction losses are significantly lower. IGBTs generally have much higher switching loss, but the use of ZVS and/or ZCS should help combat this. Whether this works well in practice remains to be seen, but preliminary tests look very promising.

The IGBTs are driven by Silabs SI8234 isolated 4A drivers. These drivers are extremely practical, with internal dead-time generation and UVLO on both gate drive outputs. As opposed to the classic high-side drivers like the IR2110, both drive outputs are isolated, and completely isolated from the input side. This allows the control electronics to be completely isolated from the mains side stuff, and unlike the classic high side drivers, it is completely immune to damage from the bridge midpoint swinging below ground, and bridge failures are not likely to kill the low voltage control electronics. Power to the upper driver is provided through a single bootstrap diode.

A classic problem with voltage fed series resonant IHs is that the power draw is maximum when unloaded, and lower when heavily loaded. Some designs are made to tolerate this condition. The disadvantage to this is that power transfer to a load will always be lower than the idle draw, and the tank VARs are badly utilized. Some designs use detuning to keep power in control. This works very well, but ZCS is lost, which can be a disadvantage when using IGBTs, especially at higher operating frequencies.

Power control is achieved through pulse skipping. As opposed to detuning, this allows ZVS to be maintained while reducing power. A second advantage to this method is that the effective switching frequency is reduced when it's skipping pulses, further lowering switching losses. The pulse skipping is implemented by sensing the tank voltage with a comparator, which triggers a flip flop that ensures that only whole cycles are skipped. As there is a 90 degree phase shift between the bridge output and the capacitor voltage, the flip flop will sample the comparator output around the peak of the tank circuit voltage. The idea of using pulse skipping is taken from an excellent paper I found

. Of course this is what Steve Conner has been doing all along in his DRSSTC driver. Just like many times before, I could have saved myself a lot of work if I used his excellent design in the first place.

One potential problem with this power control scheme is that it reacts very fast. This is generally a good thing, but when running from unsmoothed rectified mains it leads to a big problem. Since the loop is so fast, it will try to maintain a set voltage on the capacitor, so it will draw more current as the mains voltage drops. This will lead to horrible distortion of the mains current waveform and correspondingly a very bad power factor. The common solution is to slow down the feedback loop so that it rejects the mains frequency. Since the pulse skipping method needs to be fast to work properly, this is not an option here. There simplest solution to this problem is to take the setpoint for the pulse skipping comparator as a fraction of the DC bus voltage. This way, the current draw from mains will track the voltage, ensuring unity power factor.

One of the main questions I wanted to answer with this prototype was if it is possible to maintain ZVS and ZCS under all loading conditions with a simple PLL. So far it looks like the answer is no. When the phase angle is tuned for ZVS at light loading, ZVS is lost at heavy loading. If the switching phase angle is tuned for ZCS, it keeps switching at zero current independent of the loading. The ultimate goal is to minimize IGBT losses, so the best solution is anyways to tune the switching angle for lowest losses under normal operating conditions. Preliminary tests comparing ZVS and ZCS at low power (400 W input) show pretty similar IGBT losses, I will update the thread with results as I go higher in power.

Here's a schematic of the prototype. The protection circuitry is not implemented yet, and the comparator setpoint comes from the +5V rather than the DC bus.

1422919077 33 FT168388 Ih Mk2 Sch Photocopy

teravolt

Tue Feb 03 2015, 04:24AM

Registered Member #195 Joined: Fri Feb 17 2006, 08:27PM
Location: Berkeley, ca.
Posts: 1111

in your first video you have a neat frequency counter what IC are you using? do you plan to release a more formal scematic?

Wolfram

Tue Feb 03 2015, 08:43PM

Registered Member #33 Joined: Sat Feb 04 2006, 01:31PM
Location: Norway
Posts: 971

The frequency counter is based on some excellent code by Paddy Strebel

for the PIC16F84. I modified it to correct an error in the rounding algorithm, and ported it to work with my display. The code is quite clever, it uses a hybrid of frequency and period measurement for good resolution at both high and low frequencies. It's specified to work between 0.900 Hz with a resolution of 1mHz and 30MHz, though in practice it goes a bit higher depending on the prescaler of the PIC.

The schematic is up to date with the prototype, and it will be updated along with the hardware.

Wolfram

Tue Feb 10 2015, 05:50PM

Registered Member #33 Joined: Sat Feb 04 2006, 01:31PM
Location: Norway
Posts: 971

I've implemented pulse skipping for power control now. Both the lower IGBTs in the bridge are turned on for a whole cycle as long as the tank capacitor voltage exceeds the setpoint. Otherwise, the bridge outputs are driven by a 4046 PLL to maintain ZCS.

I've had loads of trouble with stability. I ran the induction heater from unsmoothed rectified mains, and set the pulse skip threshold as a DC value to cut into the peaks of the mains waveform. When the tank voltage exceeds the threshold, it starts pulse skipping but goes into some awful oscillations. The following picture shows the bridge outputs during this oscillation, at 20 microseconds per division. The DC bus voltage oscillates wildly, and actually tries to go below zero. Sorry about the crappy picture, I didn't have a USB stick handy to do a proper oscilloscope screenshot.

When running from smoothed rectified mains, the pulse skipping circuit works perfectly, and controls power smoothly from zero to full power. It also regulates instantly with load changes, as it should, and runs nicely and quietly. Here's a video showing both fullbridge outputs along with the inverter current (which is just a scaled version of the tank current due to the transformer fed series resonant tank circuit). In the beginning of the video, the tank is unloaded. A graphite crucible is put into the workcoil and removed again to show the effect of loading.

My understanding of control theory falls apart when I try to analyze where the problem is, as I'm having a hard time understanding where my phase margin exists in the circuit. I'm guessing that it's related to the time constant formed by the output impedance of my Variac and the DC bus cap, as the problem goes away when I add a lot of DC bus capacitance. Is there any way to make it more stable without adding said DC bus capacitors, or is there anything else I can do?

Kizmo

Thu Feb 12 2015, 07:36AM

Registered Member #599 Joined: Thu Mar 22 2007, 07:40PM
Location: Northern Finland, Rovaniemi
Posts: 624

I have seen something similar with my large drsstc driver based induction heater. If you start skipping pulses or interrupting the drive of the LC tank, the energy stored in the tank gets rectified back to your dc bus, have a wild guess how high the dc bus voltage will raise without any capacitance on it? :)

Uspring

Thu Feb 12 2015, 09:59AM

Registered Member #3988 Joined: Thu Jul 07 2011, 03:25PM
Location:
Posts: 711

If you clamp both sides of the full bridge, you probably won't feed much energy back into the mains, since the tank is effectively short circuited. I'm more worried about switching transients. They will lead to spikes on the mains due to mains inductance. Things get particularly bad, if you use rectified mains, since currents going into the mains don't have anywhere to go.
I don't think there's a problem with phase margins, since the circuit seems to run well if pulse skipping is derived from tank current.

Robert2

Thu Feb 12 2015, 02:45PM

Registered Member #1773 Joined: Tue Oct 21 2008, 06:56PM
Location: Poland
Posts: 93

Why do not you try with the measurement of the supply current bridge and on the basis of the value of the thyristor switch- regulate the moment that would be in the system voltage rectifier ?

I use such a system to control thyristor bridge with six three-phase mains powered.

Movie, sinusoidal- from AC power, rectangle freely adjustable :)

Wolfram

Thu Mar 05 2015, 03:54PM

Registered Member #33 Joined: Sat Feb 04 2006, 01:31PM
Location: Norway
Posts: 971

Thanks for the suggestions.

I considered the energy transfer between the tank and the bus, but I think this won't be a problem, as the bridge shorts out the load when skipping pulses.

I think mains inductance has something to do with the problem, but I can't really wrap my head around how it ends up oscillating. The problem could be the apparent negative input resistance of the bridge interacting with the bus capacitance and mains inductance. The circuit tries to maintain a constant voltage in the tank circuit, so the input power is constant, and input current sinks with rising input voltage.

In this case, the solution should be simple. Making the pulse skipping threshold track the mains voltage will invert the apparent bridge input impedance back to positive and stabilize it.

Noise doesn't seem to be a big problem so far.

I considered phase angle control with SCRs, but this will ruin the mains power factor.

Wolfram

Tue Mar 24 2015, 05:45PM

Registered Member #33 Joined: Sat Feb 04 2006, 01:31PM
Location: Norway
Posts: 971

I did end up figuring out the problem with stability, and confirmed it by fixing the problem. It's now stable under all load conditions, and I still haven't managed to break a single IGBT, or any other component for that matter.

With tank voltage control by pulse skipping, the input resistance of the bridge looks negative. As the control circuit tries to maintain a constant voltage across the tank, the power input to the bridge is constant. This means that as the bridge input voltage rises, the current draw goes down, essentially making it look like a negative resistance. This negative resistance can excite resonances in the circuit, which is exactly what happened in my case.

At first, I though the rectifier bridge would eliminate potential resonances between the DC bus capacitors and the mains inductance, like the de-Qing diode in a DC charging Tesla coil, but this is not the case. Mains keeps the diodes forward biased, and allows the DC bus cap to swing with mains inductance. I got the first clue towards this when I tried resonating my DC bus cap with the mains input. The oscillation I saw matched the frequency of the instability. I discussed this with Steve Conner, and he agreed with my interpretation of the problem, and gave me a lot more insight in the process.

To solve this problem is easy, we only need to make the bridge input appear like a positive resistance to dampen any resonance here. Making the tank capacitor voltage track the bridge DC bus voltage will accomplish this nicely, and we need to do this anyways to get reasonable mains power factor. I already tried this earlier, without any success, but I didn't understand the problem properly and made a big mistake. Bridge voltage feedback was taken through a mains transformer for isolation, and the transformer has a low pass characteristic formed by the leakage inductance, stray capacitance and load resistance. This low pass characteristic meant that it only made the bridge input resistance look positive at low frequencies, but not at the resonance frequency of the bus caps and the mains inductance.

Once I understood the problem, I tried again, but without any bandwidth limiting components in the bridge voltage sensing loop. This worked perfectly, and the waveforms are very clean and everything is stable now. I have smooth power control from almost zero up to very high powers, and IGBT losses are as low as they can be. Mains power factor is very good, and the tank current tracks mains current very cleanly without much ripples. I have full control of the switching phase angle through the PLL, and I can tune it for ZVS, ZCS and potentially both. Below is a video of it in operation, with the scope showing the operational waveforms when tuned to ZVS and close to ZCS.

Right now the prototype is limited by heating in the coupling transformer primary. I use two parallel strands of ~1mm dia. teflon insulated silver plated wire, and it gets hot enough to boil water in a few minutes even at moderate power input. The next step is to make a proper PCB for it, and implement inverter overcurrent shutdown to protect against workcoil shorts. Next will be some experiments with using RF powder cores as matching inductors for LCLR. I also have some theory written up that I'll post in this thread when time allows.

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